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Data Sheet No. PD94649
IRU3039
SYNCHRONOUS PWM CONTROLLER WITH OVER CURRENT PROTECTION FEATURES
Current Limit using Lower MOSFET Sensing Using the 6V internal regulator for charge pump circuit allows single supply operation up to 18V Programmable Switching Frequency up to 400KHz Soft-Start Function 0.8V Precision Reference Voltage Available Uncommitted Error Amplifier Available for DDR Voltage Tracking Applications Stable with Ceramic Capacitor
DESCRIPTION
The IRU3039 controller IC is designed to provide a synchronous Buck regulator and is targeted for applications where the cost and size is critical. The IRU3039 operates with a single input supply up to 18V. The output voltage can be programmed as low as 0.8V for low voltage applications. Selectable current limit is provided to tailor to external MOSFET's on-resistance for optimum cost and performance. The IRU3039 features an uncommitted error amplifier for tracking output voltage and is capable of sourcing or sinking current for applications such as DDR bus termination. This device features a programmable switching frequency set from 200KHz to 400KHz, under-voltage lockout for both Vcc and Vc supplies, an external programmable soft-start function as well as output under-voltage detection that latches off the device when an output short is detected.
APPLICATIONS
DDR Memory VDDQ/VTT Applications Graphic Card Hard Disk Drive Netcom on-board DC to DC regulator application Output voltage as low as 0.8V Low Cost On-Board DC to DC
TYPICAL APPLICATION
18V D1 C3 1uF C9 1uF L1 1uH C2 3x 15uF 25V C4 1uF C5 0.1uF Q1 IRF7466 D2 R2 L2 4.7uH Q2 IRF7458 R3 3.3V @ 8A C1 15uF
Vcc SS / SD
C7 0.1uF
VOUT2 Vc
U1 HDrv IRU3039
VP VREF Rt Comp OCSet
5.76K
LDrv
C6 2x 330uF 40mV
C11 Optional
C8 5600pF R1 14K
Gnd
PGnd
Fb
R4 1K
3.16K
Figure 1 - Typical application of IRU3039.
PACKAGE ORDER INFORMATION
TA (C) 0 To 70
Rev. 1.0 06/06/03
DEVICE IRU3039CH
PACKAGE 20-Pin MLPQ 5x5 (H)
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ABSOLUTE MAXIMUM RATINGS
Vcc Supply Voltage .................................................. -0.5V To 25V Vc Supply Voltage .................................................... -0.5V To 25V Storage Temperature Range ...................................... -65C To 150C Operating Junction Temperature Range ..................... 0C To 125C
CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device.
PACKAGE INFORMATION
20-Pin MLPQ 5x5 (H)
Co mp SS /S D OC Se t
20 VREF NC NC 1 2 3
19
18
17
16 15 VOUT2
*uJA=378C/W uJC=2.38C/W
VP
Fb
14 NC Pad 13 NC 12 11 6 7 8 9 10 Rt NC
Vcc 4 NC 5
*Exposed pad on underside is connected to a typical 1" square copper pad through vias for 4layer PCB board design.
PG nd Gn d
ELECTRICAL SPECIFICATIONS
Unless otherwise specified, specifications apply over Vcc=5V, Vc=12V and TA=0-70C. Typical values refer to 258C. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient temp. PARAMETER Feedback Voltage Fb Voltage Initial Accuracy Fb Voltage Line Regulation Reference Voltage Ref Voltage Initial Accuracy Drive Current UVLO UVLO Threshold - Vcc UVLO Hysteresis - Vcc UVLO Threshold - Vc UVLO Hysteresis - Vc UVLO Threshold - Fb Supply Current Vcc Dynamic Supply Current Vc Dynamic Supply Current Vcc Static Supply Current Vc Static Supply Current SYM VFB LREG VREF IREF TEST CONDITION MIN 0.784 4.75VRev. 1.0 06/06/03
HD rv
LD rv
Vc
0.816
UVLO VCC Supply Ramping Up UVLO VC UVLO Fb Dyn ICC Dyn IC ICCQ ICQ Supply Ramping Up Fb Ramping Down Freq=200KHz, CL=1500pF Freq=200KHz, CL=1500pF SS=0V SS=0V www.irf.com 0.3
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PARAMETER Error Amp Fb Voltage Input Bias Current Fb Voltage Input Bias Current Transconductance VP Voltage Range Soft-Start Section Charge Current Oscillator Section Frequency Ramp Amplitude Output Drivers Lo Drive Rise Time Hi Drive Rise Time Lo Drive Fall Time Hi Drive Fall Time Dead Band Time Max Duty Cycle Min Duty Cycle Internal Regulator Output Voltage Drive Current Current Limit OC Threshold Set Current OC Comp Off-Set Voltage SYM IFB1 IFB2 VP SS IB Freq VRAMP Tr(LO) Tr(HI) Tf(LO) Tf(HI) TDB DMAX DMIN VOUT2 IOUT2 IOCSET VOC(OFFSET) TEST CONDITION SS=3V SS=0V Note 1 SS=0V Rt=Open Rt=Gnd Note 1 CLOAD=1500pF, VCC=12V CLOAD=1500pF, VCC=12V CLOAD=1500pF CLOAD=1500pF HDrv going Hi or Low Fb=0.6V, Freq=200KHz Fb=1.0V Vcc=12V 5.7 40 21 -2 MIN -1 30 0.8 14 22 200 400 1.25 40 40 40 40 100 88 TYP +0.08 55 700 MAX UNITS +1 70 1.5 35 mA mA mmho V mA KHz VPP 100 100 100 100 ns ns ns ns ns % % V mA mA mV
0 6 65 28 1.5 6.3
35 5
Note 1: Guaranteed by design but not tested for production.
PIN DESCRIPTIONS
PIN# 1 4 PIN SYMBOL PIN DESCRIPTION VREF Reference Voltage. This pin can source current about 2mA. Vcc This pin provides biasing for the internal blocks of the IC as well as power for the low side FET driver. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. LDrv Output driver for the synchronous power MOSFET. PGnd This pin serves as the separate ground for MOSFET's driver and should be connected to system's ground plane. Gnd This pin serves as analog ground for internal reference and control circuitry. A high frequency capacitor must be connected from Vcc pin to this pin for noise free operation. HDrv Output driver for the high side power MOSFET. This pin should not go negative (below ground), this may cause problem for the gate drive circuit. It can happen when the inductor current goes negative (Source/Sink), soft-start at no load and for the fast load transient from full load to no load. To prevent negative voltage at gate drive, a low forward voltage drop diode might be connected between this pin and ground. Vc This pin is connected to a voltage that must be at least 4V higher than the bus voltage of the switcher (assuming 5V threshold MOSFET) and powers the high side output driver. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. Rt The switching frequency can be Programmed between 200KHz and 400KHz by connecting a resistor between Rt and Gnd. By floating the pin, the switching frequency will be 200KHz and by grounding the pin, the switching frequency will be 400KHz. www.irf.com
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PIN DESCRIPTIONS
PIN# 15 16 PIN SYMBOL PIN DESCRIPTION VOUT2 Output of internal regulator. The output is protected for short circuit. A high frequency capacitor is recommended to be connected from this pin to ground. OCSet This pin is connected to the Drain of the lower MOSFET via an external resister and it provides the positive sensing for the internal current sensing circuitry. The external resistor programs the current limit threshold depending on the RDS(ON) of the power MOSFET. An external capacitor can be placed in parallel with the programming resistor to provide high frequency noise filtering. This pin provides soft-start for the switching regulator. An internal current source charges SS / SD an external capacitor that is connected from this pin to ground which ramps up the output of the switching regulator, preventing it from overshooting as well as limiting the input current. The converter can be shutdown by pulling this pin down below 0.4V. Comp Compensation pin of the error amplifier. An external resistor and capacitor network is typically connected from this pin to ground to provide loop compensation. Fb This pin is connected directly to the output of the switching regulator via resistor divider to provide feedback to the Error amplifier. VP Non-inverting input of error amplifier. NC No connection.
17
18 19 20 2,3,5, 11,13,14
BLOCK DIAGRAM
Vcc 4 0.8V Regulator 6V 15 VOUT2
VREF 1
1.25V 0.2V
Bias Generator
3V 1.25V
4V 3V Vc 22uA 3.5V SS / SD 17 64uA Max Rt Oscillator Ct POR Error Comp 25K VP 20 25K Error Amp Enbl 0.2V
POR
12 Rt
10 Vc
9 HDrv S Q Vcc R Reset Dom 6 LDrv
Fb 19 Comp 18
3V 28uA OCSet 16 OC Comp
0.4V
FbLo Comp
7 PGnd
POR
8 Gnd
Figure 2 - Simplified block diagram of the IRU3039.
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THEORY OF OPERATION
Introduction The IRU3039 is a fixed frequency, voltage mode synchronous controller and consists of a precision reference voltage, an uncommitted error amplifier, an internal oscillator, a PWM comparator, an internal regulator, a comparator for current limit, gate drivers, soft-start and shutdown circuits (see Block Diagram). The output voltage of the synchronous converter is set and controlled by the output of the error amplifier; this is the amplified error signal from the sensed output voltage and the voltage on non-inverting input of error amplifier(VP). This voltage is compared to a fixed frequency linear sawtooth ramp and generates fixed frequency pulses of variable duty-cycle, which drives the two N-channel external MOSFETs. The timing of the IC is provided through an internal oscillator circuit which uses on-chip capacitor. The oscillation frequency is programmable between 200KHz to 400KHz by using an external resistor. Figure 14 shows switching frequency vs. external resistor (Rt). Soft-Start The IRU3039 has a programmable soft-start to control the output voltage rise and limit the current surge at the start-up. To ensure correct start-up, the soft-start sequence initiates when the Vc and Vcc rise above their threshold (3.4V and 4.4V respectively) and generates the Power On Reset (POR) signal. Soft-start function operates by sourcing an internal current to charge an external capacitor to about 3V. Initially, the soft-start function clamps the E/A's output of the PWM converter and disables the short circuit protection. During the power up, the output starts at zero and voltage at Fb is below 0.4V. The feedback UVLO is disabled during this time by injecting a current (64mA) into the Fb. This generates a voltage about 1.6V (64mA325K) across the negative input of E/A and positive input of the feedback UVLO comparator (see Figure 3).
20uA 3V SS/SD 64uA Max HDrv
POR Comp 25K 0.8V Error Amp LDrv
25K Fb
0.4V
6 4 u A325K=1.6V When SS=0
POR Feeback UVLO Comp
Figure 3 - Soft-start circuit for IRU3039. The magnitude of this current is inversely proportional to the voltage at soft-start pin. The 20mA current source starts to charge up the external capacitor. In the mean time, the soft-start voltage ramps up, the current flowing into Fb pin starts to decrease linearly and so does the voltage at the positive pin of feedback UVLO comparator and the voltage negative input of E/A. When the soft-start capacitor is around 1V, the current flowing into the Fb pin is approximately 32mA. The voltage at the positive input of the E/A is approximately: 32mA325K = 0.8V The E/A will start to operate and the output voltage starts to increase. As the soft-start capacitor voltage continues to go up, the current flowing into the Fb pin will keep decreasing. Because the voltage at pin of E/A is regulated to reference voltage 0.8V, the voltage at the Fb is: VFB = 0.8-25K3(Injected Current)
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The feedback voltage increases linearly as the injecting current goes down. The injecting current drops to zero when soft-start voltage is around 2V and the output voltage goes into steady state. As shown in Figure 4, the positive pin of feedback UVLO comparator is always higher than 0.4V, therefore, feedback UVLO is not functional during soft-start.
Output of UVLO POR 3V
Shutdown The converter can be shutdown by pulling the soft-start pin below 0.4V. The control MOSFET turns off and the synchronous MOSFET turns on during shutdown. Over-Current Protection Over-current protection is achieved with a cycle by cycle scheme and it is performed by sensing current through the RDS(ON) of low side MOSFET. As shown in Figure 5, an external resistor (RSET) is connected between OCSet pin and the drain of low side MOSFET (Q2) and sets the current limit set point. The internal current source develops a voltage across RSET. When the low side switch is turned on, the inductor current flows through the Q2 and results a voltage which is given by: VOCSET = IOCSET3RSET-RDS(ON)3iL ---(1)
2V 1V
Soft-Start Voltage Current flowing into Fb pin 0V 64uA 0uA
Voltage at negative input of Error Amp and Feedback UVLO comparator
1.6V
0.8V
IOCSET
IRU3039
OCSet RSET
Q1 L1 Q2 VOUT
0.8V
Osc
Voltage at Fb pin
0V
Figure 4 - Theoretical operational waveforms during soft-start. the output start-up time is the time period when softstart capacitor voltage increases from 1V to 2V. The startup time will be dependent on the size of the external soft-start capacitor. The start-up time can be estimated by: 20mA3TSTART/CSS = 2V-1V For a given start up time, the soft-start capacitor can be estimated as: CSS 20mA3TSTART/1V Internal Regulator The regulator powers directly from Vcc and generates a regulated voltage (6V @ 40mA). The output is protected for short circuit. This voltage can be used for charge pump circuitry as shown in Figure 1. Supply Voltage Under-Voltage Lockout The under-voltage lockout circuit assures that the MOSFET driver outputs remain in the off state whenever the supply voltage drops below set parameters. Lockout occurs if Vc or Vcc fall below 3.4V and 4.4V respectively. Normal operation resumes once Vc and Vcc rise above the set values.
Figure 5 - Diagram of the over current sensing. When voltage VOCSET is below zero, the current sensing comparator flips and disables the oscillator. The high side MOSFET is turned off and the low side MOSFET is turned on until the inductor current reduces to below current set value. The critical inductor current can be calculated by setting: VOCSET = IOCSET3RSET - RDS(ON)3IL = 0 ISET = IL(CRITICAL) = RSET3IOCSET RDS(ON) ---(2)
If the over-current condition is temporary and goes away quickly, the IRU3039 will resume its normal operation. If output is shorted or over-current condition persists, the output voltage will keep going down until it is below 0.4V. Then the output under-voltage lock out comparator goes high and turns off both MOSFETs. The operation waveforms are shown in Figure 6.
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Feedback VREF voltage 0.4V
From Figure 7, the average inductor current during the current limit mode is: IO(LIM) = ISET +
FS(NOM) IOUT
DIPK-PK(LIM) 2
---(4)
Switching frequency
The inductor's ripple current can be expressed as:
IOUT DMAX/FS(NOM)
DIPK-PK(LIM) =
(VIN - VOUT)3VOUT VIN3L3fS
High Side MOSFET turn on time (tON)
VOUT FS(NOM)3VIN
Combination of above equation and (4) results in:
IOUT
ISET = IO(LIM) -
((VIN-VOUT)3VOUT) 23fS3L3VIN [ (
---(5)
=IOUT Average Inductor Current IO(LIM) Normal operation IO(MAX) IOUT
Combination of equations (5) and (2) results in the relationship between RSET and output current limit.
Over Current Shutdown Limit Mode by UVLO
RSET =
RDS(ON) (VIN-VOUT)3VOUT 3 IO(LIM) IOCSET 23fS3L3VIN
)] ---(6)
Figure 6 - Diagram of over-current operation. Operation in current limit is shown in Figure 7, the high side MOSFET is turned off and inductor current starts to decrease. Because the output inductor current is higher than the current limit setpoint (ISET), the over-current comparator keeps high until the inductor current decreases to be below ISET. Then another cycle starts. During over-current mode, the valley inductor current is: iL(VALLEY) = ISET The peak inductor current is given as: IL(PEAK) = ISET+(VIN-VOUT)3tON/L ---(3)
Where: IO(LIM) = The Output Current Limit. Typical is 50% higher than nominal output current VIN = Maximum Input Voltage VOUT = Output Voltage fS = Switching Frequency L = Output Inductor RDS(ON) = RDS(ON) of Low Side MOSFET IOCSET = OC Threshold Set Current
To avoid undesirable trigger of over-current protection, this relationship must be satisfied: ISET / IO(NOM) DIPK-PK(NOM) 2
From the above analysis, the current limit is not only dependent on the current setting resistor RSET and RDS(ON) of low side MOSFET but it is also dependent on the input voltage, output voltage, inductance and switching frequency as well. The cycle-by-cycle over-current limit will hold for a certain amount of time, until the output voltage drops below 0.4V, the under-voltage lock out activates and latches off the output driver. The operation waveform is shown in Figure 7. Normal operation will resume after IRU3039 is powered up again.
Inductor Current
iL(PEAK) iL(AVG)
ISET =iL(VALLEY)
Current Limit Comparator Output HDrv
tON tOFF
Figure 7 - Operation waveforms during current limit.
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APPLICATION INFORMATION
Design Example: The following example is a typical application for IRU3039, the schematic is Figure 17 on page 16. VIN = 18V VOUT = 3.3V IOUT = 8A DVOUT = 100mV (output voltage ripple 3% of VOUT) fS = 200KHz Output Voltage Programming Output voltage is programmed by reference voltage and external voltage divider. The Fb pin is the inverting input of the error amplifier, which is referenced to the voltage on non-inverting pin of error amplifier. For this application, this pin (VP) is connected to reference voltage (VREF). The output voltage is defined by using the following equation: R6 VOUT = VP 3 1 + ---(7) R5 Css 203tSTART (mF) Where ---(8)
tSTART is the desired start-up time (ms)
For a start-up time of 5ms, the soft-start capacitor will be 0.1mF. Choose a ceramic capacitor at 0.1mF. Boost Supply Vc To drive the high side switch, it is necessary to supply a gate voltage at least 4V grater than the bus voltage. This is achieved by using a charge pump configuration as shown in Figure 9. This method is simple and inexpensive. The operation of the circuit is as follows: when the lower MOSFET is turned on, the capacitor (C1) is pulled down to ground and charges, up to VOUT2 value, through the diode (D1). The bus voltage will be added to this voltage when upper MOSFET turns on in next cycle, and providing supply voltage (Vc) through diode (D2). Vc is approximately: Vc VOUT2 + VBUS - (VD1 + VD2) Capacitors in the range of 0.1mF and 1mF are generally adequate for most applications. The diode must be a fast recovery device to minimize the amount of charge fed back from the charge pump capacitor into VOUT2. The diodes need to be able to block the full power rail voltage, which is seen when the high side MOSFET is switched on. For low voltage application, schottky diodes can be used to minimize forward drop across the diodes at start up.
D1
(
)
VP = VREF = 0.8V When an external resistor divider is connected to the output as shown in Figure 8.
VOUT
IRU3039
VREF VP Fb
R6 R5
Figure 8 - Typical application of the IRU3039 for programming the output voltage. Equation (7) can be rewritten as: R6 = R5 3
C3 VOUT2 Vc Regulator C2 C1 D2 VBUS Q1 L2
(
VOUT -1 VP
)
IRU3039
HDrv
Choose R5 = 1K This will result to R6 = 3.16K If the high value feedback resistors are used, the input bias current of the Fb pin could cause a slight increase in output voltage. The output voltage set point can be more accurate by using precision resistor. Soft-Start Programming The soft-start timing can be programmed by selecting the soft-start capacitance value. The start-up time of the converter can be calculated by using:
Q2
Figure 9 - Charge pump circuit. Input Capacitor Selection The input filter capacitor should be based on how much ripple the supply can tolerate on the DC input line. The ripple current generated during the on time of upper MOSFET should be provided by input capacitor. The RMS value of this ripple is expressed by:
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IRMS = IOUT D3(1-D) ---(9) Where: D is the Duty Cycle, D=VOUT/VIN. IRMS is the RMS value of the input capacitor current. IOUT is the output current for each channel. For VIN=20V, IOUT=8A and D=0.165, the IRMS=3A For higher efficiency, a low ESR capacitor is recommended. Choose three Poscap from Sanyo 25TQC15M (25V, 15mF, 90mV) with a maximum allowable ripple current of 3A. Inductor Selection The inductor is selected based on operating frequency, transient performance and allowable output voltage ripple. Low inductor value results to faster response to step load (high Di/Dt) and smaller size but will cause larger output ripple due to increase of inductor ripple current. As a rule of thumb, select an inductor that produces a ripple current of 10-40% of full load DC. For the buck converter, the inductor value for desired operating ripple current can be determined using the following relation: Di 1 VOUT VIN - VOUT = L3 ; Dt = D3 ;D= Dt fS VIN VOUT L = (VIN - VOUT)3 ---(11) VIN3Di3fS Where: VIN = Maximum Input Voltage VOUT = Output Voltage i = Inductor Ripple Current fS = Switching Frequency t = Turn On Time D = Duty Cycle If Di = 37%(IO), then the output inductor will be: L = 4.65mH The Coilcraft DO5022HC series provides a range of inductors in different values, low profile suitable for large currents, 4.7mH, 13A is a good choice for this application. This will result to a ripple approximately 37% of output current. Output Capacitor Selection The criteria to select the output capacitor is normally based on the value of the Effective Series Resistance (ESR). In general, the output capacitor must have low enough ESR to meet output ripple and load transient requirements, yet have high enough ESR to satisfy stability requirements. The ESR of the output capacitor is calculated by the following relationship: ESR [ DVO DIO ---(10)
Where: DVO = Output Voltage Ripple Di = Inductor Ripple Current DVO = 100mV and DI 40% of 8A = 3.2A This results to: ESR=31mV The Sanyo TPC series, Poscap capacitor is a good choice. The 6TPC330M, 330mF, 6.3V has an ESR 40mV. Selecting two of these capacitors in parallel, results to an ESR of 20mV which achieves our low ESR goal. The capacitor value must be high enough to absorb the inductor's ripple current. The larger the value of capacitor, the lower will be the output ripple voltage. Power MOSFET Selection The IRU3039 uses two N-Channel MOSFETs. The selections criteria to meet power transfer requirements is based on maximum drain-source voltage (VDSS), gatesource drive voltage (VGS), maximum output current, Onresistance RDS(ON) and thermal management. The MOSFET must have a maximum operating voltage (VDSS) exceeding the maximum input voltage (VIN). The gate drive requirement is almost the same for both MOSFETs. Logic-level transistor can be used and caution should be taken with devices at very low VGS to prevent undesired turn-on of the complementary MOSFET, which results a shoot-through current. The total power dissipation for MOSFETs includes conduction and switching losses. For the Buck converter, the average inductor current is equal to the DC load current. The conduction loss is defined as: PCOND(Upper Switch) = ILOAD3RDS(ON)3D3q PCOND(Lower Switch) = ILOAD3RDS(ON)3(1 - D)3q q = RDS(ON) Temperature Dependency The RDS(ON) temperature dependency should be considered for the worst case operation. This is typically given in the MOSFET data sheet. Ensure that the conduction losses and switching losses do not exceed the package ratings or violate the overall thermal budget.
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Choose IRF7466 for control MOSFET and IRF7458 for synchronous MOSFET. These devices provide low onresistance in a compact SOIC 8-Pin package. The MOSFETs have the following data: IRF7466 VDSS = 30V ID = 11A RDS(ON) = 12.5mV IRF7458 VDSS = 30V ID = 14A RDS(ON) = 8mV These values are taken under a certain condition test. For more details please refer to the IRF7466 and IRF7458 data sheets. By using equation (12), we can calculate the total switching losses. PSW(TOTAL) = 92mW Programming the Over-Current Limit The over-current threshold can be set by connecting a resistor (RSET) from drain of low side MOSFET to the OCSet pin. The resistor can be calculated by using equation (2). The RDS(ON) has a positive temperature coefficient and it should be considered for the worse case operation. RDS(ON) = 8mV31.5 = 12mV ISET IO(LIM) = 8A31.5 = 12A (50% over nominal output current) This results to: RSET = 5.76KV Feedback Compensation The IRU3039 is a voltage mode controller; the control loop is a single voltage feedback path including error amplifier and error comparator. To achieve fast transient response and accurate output regulation, a compensation circuit is necessary. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency and adequate phase margin (greater than 458). The output LC filter introduces a double pole, -40dB/ decade gain slope above its corner resonant frequency, and a total phase lag of 1808 (see Figure 11). The Resonant frequency of the LC filter is expressed as follows: FLC = 1 2p3 LO3CO ---(13)
The total conduction losses will be: PCON(TOTAL) = PCON(UPPER) + PCON(LOWER) PCON(TOTAL) = 0.85W The switching loss is more difficult to calculate, even though the switching transition is well understood. The reason is the effect of the parasitic components and switching times during the switching procedures such as turn-on / turnoff delays and rise and fall times. The control MOSFET contributes to the majority of the switching losses in synchronous Buck converter. The synchronous MOSFET turns on under zero voltage conditions, therefore, the turn on losses for synchronous MOSFET can be neglected. With a linear approximation, the total switching loss can be expressed as: VDS(OFF) tr + tf 3 3 ILOAD ---(12) 2 T Where: VDS(OFF) = Drain to Source Voltage at off time tr = Rise Time tf = Fall Time T = Switching Period ILOAD = Load Current PSW = The switching time waveform is shown in Figure 10.
VDS 90%
Figure 11 shows gain and phase of the LC filter. Since we already have 1808 phase shift just from the output filter, the system risks being unstable.
10% VGS td(ON) tr td(OFF) tf
Gain 0dB -40dB/decade Phase 08
Figure 10 - Switching time waveforms. From IRF7466 data sheet we obtain: IRF7466
-1808 FLC Frequency FLC Frequency
tr = 2.8ns tf = 3.6ns
Figure 11 - Gain and phase of LC filter.
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The IRU3039's error amplifier is a differential-input transconductance amplifier. The output is available for DC gain control or AC phase compensation. The E/A can be compensated with or without the use of local feedback. When operated without local feedback, the transconductance properties of the E/A become evident and can be used to cancel one of the output filter poles. This will be accomplished with a series RC circuit from Comp pin to ground as shown in Figure 12. Note that this method requires that the output capacitor should have enough ESR to satisfy stability requirements. In general, the output capacitor's ESR generates a zero typically at 5KHz to 50KHz which is essential for an acceptable phase margin. The ESR zero of the output capacitor expressed as follows: 1 FESR = ---(14) 2p3ESR3Co
VOUT R6 Fb
First select the desired zero-crossover frequency (Fo): Fo > FESR and FO [ (1/5 ~ 1/10)3fS Use the following equation to calculate R4: R4 = 1 VOSC Fo3FESR R5 + R6 3 3 3 gm VIN FLC2 R5 ---(18)
Where: VIN = Maximum Input Voltage VOSC = Oscillator Ramp Voltage Fo = Crossover Frequency FESR = Zero Frequency of the Output Capacitor FLC = Resonant Frequency of the Output Filter R5 and R6 = Resistor Dividers for Output Voltage Programming gm = Error Amplifier Transconductance For: VIN = 18V VOSC = 3.3V Fo = 20KHz FESR = 12KHz FLC = 2.8KHz R5 = 1K R6 = 3.16K gm = 700mmho
This results to R4=12.08K Choose R4=14K
R5 Vp=VREF
Gain(dB)
E/A
Comp V e C9 R4
To cancel one of the LC filter poles, place the zero before the LC filter resonant frequency pole: FZ 75%FLC FZ 0.753 1 2p LO 3 CO ---(19)
H(s) dB
For: Lo = 4.7mH Co = 660mF
FZ
Frequency
FZ = 2.1KHz R4 = 14K
Using equations (17) and (19) to calculate C9, we get: C9 5300pF; Choose C9 =5600pF One more capacitor is sometimes added in parallel with C9 and R4. This introduces one more pole which is mainly used to suppress the switching noise. The additional pole is given by: FP = 1 2p3R43 C93CPOLE C9 + CPOLE
Figure 12 - Compensation network without local feedback and its asymptotic gain plot. The transfer function (Ve / VOUT) is given by: H(s) = gm3
(
R5 1 + sR4C9 3 R6 + R5 sC9
)
---(15)
The (s) indicates that the transfer function varies as a function of frequency. This configuration introduces a gain and zero, expressed by: |H(s=j32p3FO)| = gm3 FZ = 1 2p3R43C9 R5 3R4 R63R5 ---(17) ---(16)
The pole sets to one half of switching frequency which results in the capacitor CPOLE: CPOLE = 1 p3R43fS - 1 C9 fS for FP << 2 1 p3R43fS
|H(s)| is the gain at zero cross frequency.
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For a general solution for unconditionally stability for ceramic capacitor with very low ESR and any type of output capacitors, in a wide range of ESR values we should implement local feedback with a compensation network. The typically used compensation network for voltage-mode controller is shown in Figure 13.
ZIN C10 R8 R6 Fb R5 Vp=V REF
Gain(dB)
FP1 = 0 FP2 = FP3 = 1 2p3R83C10 1 2p3R73 FZ1 =
( C123C11) C12+C11
1 2p3R73C12
VOUT
C12 R7 C11 Zf
1 2p3R73C11
1 1 FZ2 = 2p3C103(R6 + R8) 2p3C103R6 Cross Over Frequency:
E/A
Comp
V e
FO = R73C103
VIN 1 3 VOSC 2p3Lo3Co
---(21)
H(s) dB
Where: VIN = Maximum Input Voltage VOSC = Oscillator Ramp Voltage Lo = Output Inductor Co = Total Output Capacitors The stability requirement will be satisfied by placing the poles and zeros of the compensation network according to following design rules. The consideration has been taken to satisfy condition (20) regarding transconductance error amplifier. These design rules will give a crossover frequency approximately one-tenth of the switching frequency. The higher the band width, the potentially faster the load transient speed. The gain margin will be large enough to provide high DC-regulation accuracy (typically -5dB to 12dB). The phase margin should be greater than 458 for overall stability. Based on the frequency of the zero generated by ESR versus crossover frequency, the compensation type can be different. The table below shows the compensation type and location of crossover frequency. Compensator Location of Zero Typical Type Crossover Frequency Output (FO) Capacitor Type II (PI) FPO < FZO < FO < fS/2 Electrolytic, Tantalum Type III (PID) FPO < FO < FZO < fS/2 Tantalum, Method A Ceramic Type III (PID) FPO < FO < fS/2 < FZO Ceramic Method B Table - The compensation type and location of zero crossover frequency. Detail information is dicussed in application Note AN1043 which can be downloaded from the IR Web-Site.
FZ1
FZ2
FP2
FP3
Frequency
Figure 13 - Compensation network with local feedback and its asymptotic gain plot. In such configuration, the transfer function is given by: Ve 1 - gmZf = VOUT 1 + gmZIN The error amplifier gain is independent of the transconductance under the following condition: gmZf >> 1 and gmZIN >>1 ---(20)
By replacing ZIN and Zf according to Figure 9, the transformer function can be expressed as: H(s) = (1+sR7C11)3[1+sC10(R6+R8)] 1 3 sR6(C12+C11) C12C11 1+sR7 C12+C11 3(1+sR8C10)
[
(
)]
As known, transconductance amplifier has high impedance (current source) output, therefore, consider should be taken when loading the E/A output. It may exceed its source/sink output current capability, so that the amplifier will not be able to swing its output voltage over the necessary range. The compensation network has three poles and two zeros and they are expressed as follows:
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IRU3039
Layout Consideration The layout is very important when designing high frequency switching converters. Layout will affect noise pickup and can cause a good design to perform with less than expected results. Start to place the power components. Make all the connections in the top layer with wide, copper filled areas. The inductor, output capacitor and the MOSFET should be close to each other as possible. This helps to reduce the EMI radiated by the power traces due to the high switching currents through them. Place input capacitor directly to the drain of the high-side MOSFET. To reduce the ESR, replace the single input capacitor with two parallel units. The feedback part of the system should be kept away from the inductor and other noise sources and be placed close to the IC. In multilayer PCB, use one layer as power ground plane and have a separate control circuit ground (analog ground), to which all signals are referenced. The goal is to localize the high current path to a separate loop that does not interfere with the more sensitive analog control function. These two grounds must be connected together on the PC board layout at a single point.
450
400
Frequency (KHz)
350
300
250
200 0 50 100 150 200 V Rt (KV ) 250 300 350 400 450
Figure 14 - Switching Frequency versus Resistor.
Rev. 1.0 06/06/03
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13
IRU3039
TYPICAL APPLICATION
5V +12V C3 1uF Vcc SS / SD C7 0.1uF VP VREF Rt Comp Gnd Vc HDrv D1 R2 7.12K LDrv VOUT2 R3 PGnd Fb R4 1K 2.15K Q2 IRF7457 C4 1uF L1 1uH C2 2x 150uF C1 47uF
U1 IRU3039
OCSet
Q1 IRF7457 L2 3.3uH VOUT 2.5V @ 10A C6 2x 330uF, 40mV
C8 2200pF R1 28K
Figure 15 - Typical application of the IRU3039 with two input supplies.
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Rev. 1.0 06/06/03
IRU3039
TYPICAL APPLICATION
5V
C1 1uF
12V
C2 1uF
L1 1uH C3 2x 100uF, 55mV 10TPB100M
5V
C4 47uF
Vcc VREF VP
Vc VOUT2
Q1 1/2 of IRF7313 D1 R7 8K Q1 1/2 of IRF7313 L2 4.7uH
U1 HDrv IRU3039
OCSet LDrv
C6 0.1uF
SS / SD
Rt Comp
C8 4.7nF R2 27K
PGnd
R1
C7 2x 330uF, 40mV 6TPC330M
VDDQ 1.8V @ 5A
Fb Gnd
1.25K R3 1K
5V
C9 1uF
12V
C10 1uF C11 100uF, 55mV 10TPB100M
VREF Vcc
R4 1K
Vc
Q2 1/2 of IRF7313 D2 L3
VP
R5 1K
U2 HDrv IRU3038
LDrv
C12 0.15uF
SS / SD PGnd Fb Gnd
Q2 1/2 of IRF7313
4.7uH
VTT (0.9V @ 3A)
C13 2x 330uF, 40mV 6TPC330M
C14 5.6nF R6 13K
Rt Comp
Figure 16 - Typical application of IRU3039 for DDR memory when IRU3039 generates VCORE and IRU3038 generates the termination voltage.
Rev. 1.0 06/06/03
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IRU3039
DEMO-BOARD APPLICATION
18V to 3.3V @ 8A 18V D1 C11 1uF Vcc SS / SD C6 0.1uF C10 0.1uF C7 5600pF R7 14K VP VREF Rt Comp Gnd PGnd C3 C13 1uF 1uF VOUT2 Vc C5 0.1uF L1 1uH C2A,B,C 3x 15uF 25V Q1 IRF7466 C1 15uF
U1 HDrv IRU3039
OCSet LDrv
R4 5.76K
D2
L2 3.3V @ 8A C9A,B 2x 330uF 40mV
Q2 IRF7458
4.7uH C8 470pF C12 R8 1uF 4.7V R9
Fb R10 1K
3.16K
Figure 17 - Demo-board application of the IRU3039. PARTS LIST Ref Desig Description Q1 MOSFET Q2 MOSFET U1 Controller D1 Schottky Diode D2 Schottky Diode L1 Inductor L2 Inductor C1,C2A,B,C Cap, Poscap C5,6,10 Capacitor C7 Capacitor C8 Capacitor C9A,B Capacitor C3,11,12 Capacitor C13 Capacitor R4 Resistor R7 Resistor R8 Resistor R9 Resistor R10 Resistor Value Qty Part# 30V, 12.5mV, 11A 1 IRF7466 30V, 8mV, 14A 1 IRF7458 Synchronous PWM 1 IRU3039 Fast Switching 1 BAT54S Fast Switching 1 BAT54 1mH, 3A 1 DS1608C-102 4.7mH, 13A 1 DO5022P-472HC 15mF, 25V 4 25TQC15M 0.1mF, Y5V, 25V 3 ECJ-2VF1E104Z 5600pF, X7R, 50V 1 ECU-V1H562KBG 470pF, X7R, 50V 1 ECJ-2VC1H471J 330uF, 40mV 2 6TPB-330M 1mF, Y5V, 16V 3 ECJ-2VF1C105Z 1mF, X7R, 25V 1 ECJ-3YB1E105K 5.76K 1 14K 1 4.7V 1 3.16K 1 1K 1 Manuf IR IR IR IR IR Coilcraft Coilcraft Sanyo Panasonic Panasonic Panasonic Sanyo Panasonic Panasonic Web site (www.) irf.com
coilcraft.com sanyo.com maco.panasonic.co.jp
sanyo.com maco.panasonic.co.jp
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Rev. 1.0 06/06/03
IRU3039
TYPICAL OPERATING CHARACTERISTICS
Test Conditions: VIN=20V, VOUT=3.3V, IOUT=0-8A, Fs=200KHz
Figure 18 - Normal condition at No Load. Ch1: HDrv, Ch2: LDrv, Ch4: Inductor Current
Figure 20 - Soft-Start. Ch1: VIN, Ch2: VOUT, Ch3: VOUT2, Ch4: Vss
Figure 19 - Soft-Start pin grounded. Ch1: HDrv, Ch2: LDrv
Figure 21 - Output Ripple. Ch1: Output Ripple, Ch2: HDrv, Ch3: LDrv, Ch4: Inductor Current
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17
IRU3039
TYPICAL OPERATING CHARACTERISTICS
Test Conditions: VIN=20V, VOUT=3.3V, IOUT=0-8A, Fs=200KHz
8A
0A
Figure 22 - Output shorted at start up. Ch1: VOUT, Ch3: Vss, Ch4: Inductor Current
Figure 23 - Load Transient Response Ch1: VOUT, Ch3: Output Current
90 88 86
Efficiency (%)
84 82 80 78 76 74 72 70 0 1 2 3 4 5 6 7 8 9 10 11
Output Current (A)
Figure 24 - Efficiency Measurement. VIN=20V, VOUT=3.3V
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Rev. 1.0 06/06/03
IRU3039
TYPICAL PERFORMANCE CHARACTERISTICS
For all charts: VC=VCC=12V, 20V, 24V
Note: Data are taken with few samples to indicate the variation of these parameters over the wide temperature range.
0.802
6.2
0.801
6.15
0.8
6.1
0.799
0.798
12 Volt 20 Volt 24 Volt
Vout2 (V)
6.05 6 5.95
12 Volt 20 Volt
Vref (V)
0.797
0.796
5.9
0.795 0.794
24 Volt
5.85 5.8 -45
0.793 -45 -10 25 60 95 130
-10
25
60
95
130
Temperature (C)
Temperature (C)
Figure 25 - VREF vs.Temperature
Figure 28 - VOUT2 vs.Temperature
215 214 213 212
12 Volt
410
405
Frequency (KHz)
211
24 Volt
Frequency (KHz)
20 Volt
12 Volt
400
20 Volt 24 Volt
210 209 208 207 206 205 204 -45 -10 25 60 95 130
395
390
385 -45 -10 25 60 95 130
Temperature (C)
Temperature (C)
Figure 26 - Frequency vs.Temperature FS=200KHz
Figure 29 - Frequency vs.Temperature FS=400KHz
200 Deadtime, Switch FET Drive Rising Time (ns) Deadtime, Sync FET Drive Rising Time (ns)
200
150
150
100
12 Volts 20 Volts 24 Volts
100
12 Volts 20 Volts 24 Volts
50
50
0 -45 -10 25 60 Temperature (C) 95 130
0 -45 -10 25 60 Temperature (C) 95 130
Figure 27 - Deadtime, Control FET Drive Rising Time vs.Temperature FS=400KHz, CLOAD=3300pF
Rev. 1.0 06/06/03
Figure 30 - Deadtime, Sync FET Drive Rising Time vs.Temperature FS=400KHz, CLOAD=3300pF www.irf.com
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IRU3039
TYPICAL PERFORMANCE CHARACTERISTICS
For all charts: VC=VCC=12V, 20V, 24V
Note: Data are taken with few samples to indicate the variation of these parameters over the wide temperature range.
200 180 Control FET Drive Rise Time (ns) 160 140 120 100 80 60 40 20 0 -45 -10 25 60 Temperature (C) 95 130
12 Volts 20 Volts 24 Volts
200 180 Control FET Drive Fall Time (ns) 160 140 120 100 80 60 40 20 0 -45 5 55 Temperature (C) 105 155
12 Volts 20 Volts 24 Volts
Figure 31 - Control FET Drive Rise Time vs.Temp. FS=400KHz, CLOAD=3300pF
Figure 33 - Control FET Drive Fall Time vs.Temp. FS=400KHz, CLOAD=3300pF
200 180 160 Sync FET Drive Rise Time (ns) 140 120 100 80 60 40 20 0 -45 -10 25 60 Temperature (C) 95 130
12 Volts 20 Volts 24 Volts
20 18 16 Sync FET Drive Time (ns) 14 12 10 8 6 4 2 0 -45 -10 25 60 Temperature (C) 95 130
12 Volts 20 Volts 24 Volts
Figure 32 - Sync FET Drive Rise Time vs.Temp. FS=400KHz, CLOAD=3300pF
Figure 34 - Sync FET Drive Fall Time vs.Temp. FS=400KHz, CLOAD=3300pF
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 02/01
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Rev. 1.0 06/06/03
IRU3039
(H) MLPQ 5x5 Package 20-Pin
D D/2 D2 EXPOSED PAD
PIN NUMBER 1 PIN 1 MARK AREA (See Note1)
E/2
E2 E R
L
TOP VIEW
e
BOTTOM VIEW
B
A A3 SIDE VIEW A1
Note 1: Details of pin #1 are optional, but must be located within the zone indicated. The identifier may be molded, or marked features. 20-PIN 5x5
SYMBOL DESIG A A1 A3 B D D2 E E2 e L R
MIN 0.80 0.00 0.23
NOM MAX 0.90 1.00 0.02 0.05 0.20 REF
0.30 0.38 5.00 BSC 3.00 3.15 3.25 5.00 BSC 3.00 3.15 3.25 0.65 BSC 0.45 0.115 0.55 --0.65 ---
NOTE: ALL MEASUREMENTS ARE IN MILLIMETERS.
Rev. 1.0 06/06/03
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21
IRU3039
PACKAGE SHIPMENT METHOD
PKG DESIG H PACKAGE DESCRIPTION MLPQ 5x5 PIN COUNT 20 PARTS PER TUBE --PARTS PER REEL 3000 T&R Orientation Fig A
1
1
1
Feed Direction Figure A - Live Bug
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 02/01
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